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ts1002/TS1004 page 1 ? 2014 silicon laboratories, inc. all rights reserved. features single 0.65v to 2.5v operation supply current: 0.6 a per amplifier (typ) offset voltage: 0.5mv (typ) low tcv os : 10v/c (typ) a vol driving 100k ? load: 90db (min) unity gain stable rail-to-rail input and output no output phase reversal packaging: ts1002 ? 8-pin msop TS1004 ? 14-pin tssop applications battery/solar-powered instrumentation portable gas monitors low-voltage signal processing nanopower active filters wireless remote sensors battery-powered industrial sensors active rfid readers powerline or battery current sensing handheld/portable pos terminals description the ts1002 and the TS1004 are the industry?s first and only dual and quad single-supply, precision cmos operational amplifiers fully specified to operate at 0.8v while consuming less than 0.6a supply current per amplifier. optimized for ultra-long-life battery-powered applications, the ts1002 and the TS1004 join the ts1001 operati onal amplifier in the ?nanowatt analog?? high-performance analog integrated circuits portfolio. both op amps exhibit a typical offset voltage of 0.5mv, a typical input bias current of 25pa, and rail-to-rail input and output stages. the ts1002 and the TS1004 can operate from single-supply voltages from 0.65v to 2.5v. the ts1002/TS1004?s combined features make either an excellent choice in applications where very low supply current and low operating supply voltage translate into very long equipment operating time. applications include: nanopower active filters, wireless remote sensors, battery and powerline current sensors, portable gas monitors, and handheld/portable pos terminals. the ts1002 and the TS1004 are fully specified at vdd = 0.8v and over the industrial temperature range ( ? 40c to +85c). the ts1002 is available in pcb-space saving 8-lead msop surface-mount packages. the TS1004 is available in a 14-pin tssop package. the only 0.8v/0.6a rail-to-rail dual/quad op amps typical application circuit a nanowatt 2-pole sallen-key low-pass filter v dd = 0.8v 0.53 0.58 0.63 0.68 0.73 percent of units - % 30% 25% 20% 15% 10% 5% 0% supply current distribution supply current per amplifier - a
ts1002/TS1004 page 2 ts1002/4 rev. 1.0 absolute maximum ratings total supply voltage (v dd to v ss ) ........................... +2.75 v voltage inputs (in+, in-) ........... (v ss - 0.3v) to (v dd + 0.3v) differential input voltage .......................................... 2.75 v input current (in+ , in-) .............................................. 20 ma output short-circuit dura tion to gnd .................... indefinite continuous power dissipation (t a = +70c) 8-pin msop (derate 7mw/c above +70c) ...... 450 mw 14-pin tssop (derate 8.3mw/c above +70c) ............................................................................ 500 mw operating temper ature range .................... - 40c to +85c junction temper ature .............................................. +150c storage temperature rang e ..................... -65c to +150c lead temperature (soldering, 10s ) ............................. +300 electrical and thermal stresses beyond thos e listed under ?absolute maximum ratings? ma y cause permanent damage to the device. these are stress ratings only and functional operati on of the device at these or any other condition beyond those indicated in the op erational sections of the specifications is not implied. exposure to any absolute maximum rating conditions for extended periods may affect device reliability and lifetime. package/ordering information tape & reel order number part marking package quantity tape & reel order number part marking package quantity ts1002im8 tadj ----- TS1004it14 t1004i ----- ts1002im8t 2500 TS1004it14t 2500 lead-free program: silicon labs supplies only lead-free packaging. consult silicon labs for produ cts specified with wider oper ating temperature ranges.
ts1002/TS1004 ts1002/4 rev. 1.0 page 3 electrical characteristics v dd = +0.8v, v ss = 0v, v incm = v ss ; r l = 100k ? to (v dd -v ss )/2; t a = -40c to +85c, unless otherwise noted. typical values are at t a = +25c. see note 1 parameters symbol conditions min typ max units supply voltage range v dd -v ss 0.65 0.8 2.5 v supply current i sy ts1002; r l = open circuit t a = 25 c 1.2 1.6 a -40 c t a 85 c 2 TS1004; r l = open circuit t a = 25 c 2.4 3.2 a -40 c t a 85 c 4 input offset voltage v os v in = v ss or v dd t a = 25 c 0.5 3 mv -40 c t a 85 c 5 input offset voltage drift tcv os 10 v/c input bias current i in+ , i in- v in+ , v in- = (v dd - v ss )/2 t a = 25 c 0.025 na -40 c t a 85 c 20 input offset current i os specified as i in+ - i in- v in+ , v in- = (v dd - v ss )/2 t a = 25 c 0.01 na -40 c t a 85 c 2 input voltage range ivr guaranteed by input offset voltage test v ss v dd v common-mode rejection ratio cmrr 0v v in ( cm ) 0.4v 50 74 db power supply rejection ratio psrr 0.65v (v dd - v ss ) 2.5v 50 74 db output voltage high v oh specified as v dd - v out , r l = 100k ? to v ss t a = 25 c 1.2 2 mv -40 c t a 85 c 2.5 specified as v dd - v out , r l = 10k ? to v ss t a = 25 c 10 16 -40 c t a 85 c 20 output voltage low v ol specified as v out - v ss , r l = 100k ? to v dd t a = 25 c 0.4 0.6 mv -40 c t a 85 c 1 specified as v out - v ss , r l = 10k ? to v dd t a = 25 c 5 7 -40 c t a 85 c 10 short-circuit current i sc+ v out = v ss t a = 25 c 0.5 1.5 ma -40 c t a 85 c 0.3 i sc- v out = v dd t a = 25 c 4.5 11 -40 c t a 85 c 3 open-loop voltage gain a vol v ss +50mv v out v dd -50mv t a = 25 c 90 104 db -40 c t a 85 c 85 gain-bandwidth product gbwp r l = 100k ? to v ss , c l = 20pf 4 khz phase margin m unity-gain crossover, r l = 100k ? to v ss , c l = 20pf 70 degrees slew rate sr r l = 100k ? to v ss, a vcl = +1v/v 1.5 v/ms full-power bandwidth fpbw fpbw = sr/( ? v out,pp ); v out,pp = 0.7v pp 680 hz input voltage noise density e n f = 1khz 0.6 v/ note 1: all specifications are 100% tested at t a = +25c. specification lim its over temperature (t a = t min to t max ) are guaranteed by device characterization, not production tested.
ts1002/TS1004 page 4 ts1002/4 rev. 1.0 typical performance characteristics total supply current vs supply voltage supply curent - a supply voltage - volt total supply current vs input common-mode voltage supply curent - a input common-mode voltage - volt total supply current vs input common-mode voltage input offset voltage vs input common-mode voltage input offset voltage - mv input offset voltage - mv input common-mode voltage - volt input offset voltage vs supply voltage input common-mode voltage - volt supply curent - a supply voltage - volt v dd =0.8v t a = +25 c input offset voltage vs input common-mode voltage input offset voltage - mv input common-mode voltage - volt v dd = 2.5v t a = +25 c +25c, TS1004 +85c, TS1004 -40c, TS1004 0.6 1 1.5 t a = +25 c 1.56 1.34 1.12 0.9 0 0.2 0.4 0.6 0.8 t a = +25 c 0 0.5 1.5 2 2.5 1 t a = +25 c 0.5 1.5 2 2.5 1 v incm = v dd 0.65 0.6 0.55 0.5 0.55 v incm = 0v 0 0.2 0.4 0.6 0.8 1 0.5 0 -1 -0.5 1 0.5 0 -1 -0.5 0 0.5 1.5 2 2.5 1 1.11 1.58 0.65 2.5 2.04 1.9 2.4 2.8 +85c, ts1002 +25c, ts1002 -40c, ts1002 1.78 2 TS1004 ts1002 1.56 1.34 1.12 0.9 1.78 2 TS1004 ts1002
ts1002/TS1004 ts1002/4 rev. 1.0 page 5 -40 typical performance characteristics input bias current (i in+ , i in- ) vs input common-mode voltage input bias current - p a input common-mode voltage - volt output voltage high (v oh ) vs temperature, r load =100k ? temperature - c output voltage low (v ol ) vs temperature, r load =100k ? temperature - c output voltage high (v oh ) vs temperature, r load =10k ? output voltage low (v ol ) vs temperature, r load =10k ? input bias current (i in+ , i in- ) vs input common-mode voltage output saturation voltage - mv input common-mode voltage - volt input bias current - p a output saturation voltage - mv v dd = 2.5v t a = +25 c v dd =0.8v t a = +25 c 0 0.2 0.4 0.6 0.8 0 0.5 1.5 2 2.5 1 100 75 25 -50 -25 0 50 250 200 100 -50 0 50 150 r l = 100k ? v dd = 0.8v v dd = 2.5v r l = 100k ? v dd = 0.8v v dd = 2.5v 4.5 4 2 0 1 3 3.5 0.5 1.5 2.5 1.8 1.6 0.8 0 0.4 1.2 1.4 0.2 0.6 1 +25 +85 -40 +25 +85 20 0 10 30 35 5 15 25 output saturation voltage - mv output saturation voltage - mv temperature - c temperature - c -40 +25 +85 -40 +25 +85 r l = 10k ? v dd = 0.8v v dd = 2.5v r l = 10k ? v dd = 0.8v v dd = 2.5v 16 0 8 4 12 20
ts1002/TS1004 page 6 ts1002/4 rev. 1.0 v out(n) - 100v/div 0.1hz to 10hz output voltage noise typical performance characteristics output short circuit current, i sc+ vs temperature output short-circuit current - ma output short circuit current, i sc- vs temperature large-signal transient response v dd = 2.5v, v ss = gnd, r load = 100k ?, c load = 15pf 200 s/div output short-circuit current - ma input small-signal transient response v dd = 2.5v, v ss = gnd, r load = 100k ?, c load = 15pf 2ms/div output input output temperature - c temperature - c -40 +25 +85 -40 +25 +85 v dd = 0.8v v dd = 2.5v v dd = 0.8v v dd = 2.5v v out = 0v v out = v dd 25 15 0 5 10 20 50 30 0 10 20 40 60 70 gain and phase vs. frequency gain - db frequency - hz phase - de g rees 10 1k 10k 100 40 -20 0 20 60 50 -250 -150 -50 150 100k v dd = 0.8v t a = +25 c r l = 100k ? c l = 20pf a vcl = 1000 v/v phase gain 4khz 70 1 second/div 130v pp
ts1002/TS1004 ts1002/4 rev. 1.0 page 7 pin functions pin label function ts1002 msop TS1004 tssop 1, 7 1, 7, 8, 14 out amplifier output s: a, b ? ts1002; a, b, c, d ? TS1004 4 7 v ss negative supply or analog gnd. if applying a negative voltage to this pin, connect a 0.1f capacitor from this pin to analog gnd. 3, 5 3, 5, 10, 12 +in amplifier non-inverting inputs: a, b ? ts1002; a, b, c, d ? TS1004 2, 6 2, 6, 9, 13 -in amplifier inverting inputs: a, b ? ts1002; a, b, c, d ? TS1004 8 14 v dd positive supply connection. connect a 0.1f bypass capacitor from this pin to analog gnd. theory of operation the ts1002 and the TS1004 are fully functional for input signals from the negative supply (v ss or gnd) to the positive supply (v dd ). their input stages consist of two differential amplifiers, a p-channel cmos stage and an n-channel cmos stage that are active over different ranges of the input common mode voltage. the p-channel input pair is active for input common mode voltages, v incm , between the negative supply to approximately 0.4v below the positive supply. as the common-mode input voltage moves closer towards v dd , an internal current mirror activates the n-channel input pair differential pair. the p-channel input pair becomes inactive for the balance of the input common mode voltage range up to the positive supply. because both input stages have their own offset voltage (v os ) characteristic, the offset voltage of these amplifiers is a function of the applied input common-mode voltage, v incm . the v os has a crossover point at ~0.4v from v dd (refer to the v os vs. v cm curve in the typical operating characteristics section). caution should be taken in applications where the input signal amplitude is comparable to the amplifiers? v os value and/or the design requires high accuracy. in these situations, it is necessary for the input signal to avoid the crossover point. in addition, amplifier parameters such as psrr and cmrr which involve the input offset voltage will also be affected by changes in the input common-mode voltage across the differential pair transition region. the amplifiers? second stage is a folded-cascode transistor arrangement that converts the input stage differential signals into a single-ended output. a complementary drive generator supplies current to the output transistors that swing rail to rail. the amplifiers? output stage voltage swings within 1.2mv from the rails at 0.8v supply when driving an output load of 100k ? - which provides the maximum possible dynamic range at the output. this is particularly important when operating on low supply voltages. when driving a stiffer 10k ? load, the amplifiers? output swings within 10mv of v dd and within 5mv of v ss (or gnd). applications information portable gas detection sensor amplifier gas sensors are used in m any different industrial and medical applications. gas sensors generate a current that is proporti onal to the percentage of a particular gas concentration sensed in an air sample. this output current flows through a load resistor and the resultant voltage drop is amplified. depending on the sensed gas and sensitivity of the sensor, the output current can be in the range of tens of microamperes to a few milliamperes. gas sensor datasheets often specify a recommended load resistor value or a ra nge of load resistors from which to choose. there are two main applications for oxygen sensors ? applications which sense oxygen when it is abundantly present (that is, in air or near an oxygen tank) and those which detect traces of oxygen in parts-per-million concentration. in medical applications, oxygen sensors are used when air quality or oxygen delivered to a patient needs to be monitored. in fresh air, the concentration of oxygen
ts1002/TS1004 page 8 ts1002/4 rev. 1.0 is 20.9% and air samples containing less than 18% oxygen are considered dangerous. in industrial applications, oxygen sensors are used to detect the absence of oxygen; for example, vacuum-packaging of food products is one example. the circuit in figure 1 illustrates a typical implementation used to amplify the output of an oxygen detector. either amplifier makes an excellent choice for this application as it only draws 0.6a of supply current pe r amplifier and operates on supply voltages down to 0.65v. with the components shown in the figure, the circuit consumes less than 0.7 a of supply current ensuring that small form-factor single- or button-cell batteries (exhibiting low mah charge ratings) could last beyond the operating life of the oxygen sensor. the precision specifications of these amplifiers, such as their low offset voltage, low tcv os , low input bias current, high cmrr, and high psrr are other factors which make these amplifiers excellent choices for this application. since oxygen sensors typically exhibit an operating life of one to two years, an oxygen sensor amplifier built around one of these amplifiers can operate from a conventionally- available single 1.5-v alkaline aa battery for over 290 years! at such low power consumption from a single cell, the oxygen sensor could be replaced over 150 times before the battery requires replacing! nanowatt, buffered single-pole low-pass filters when receiving low-level signals, limiting the bandwidth of the incoming sign als into the system is often required. as shown in figure 2, the simplest way to achieve this objective is to use an rc filter at the noninverting terminal of the amplifier. if additional attenuation is needed, a two-pole sallen-key filter can be used to provide the additional attenuation as shown in figure 3. for best results, the filt er?s cutoff frequency should be 8 to 10 times lower than the amplfier?s crossover frequency. additional operational amplifier phase margin shift can be avoided if the amplifier bandwidth-to-signal bandwidth ratio is greater than 8. the design equations for the 2-pole sallen-key low- pass filter are given below with component values selected to set a 400hz low-pass filter cutoff frequency: r1 = r2 = r = 1m ? c1 = c2 = c = 400pf q = filter peaking factor = 1 f?3db = 1/(2 x x rc) = 400 hz r3 = r4/(2-1/q); with q = 1, r3 = r4. a single +1.5 v supply, two op amp instrumentation amplifier the amplifiers? ultra-low supply current and ultra-low voltage operation make them ideal for battery- powered applications such as the instrumentation amplifier shown in figure 4 using a ts1002. figure 2: a simple, single-pole active low-pass filter. figure 3: a nanopower 2-pole sallen-key low-pass filter. figure 1 : a nanopower, precision oxygen gas sensor amplifier.
ts1002/TS1004 ts1002/4 rev. 1.0 page 9 the circuit utilizes the classic two op amp instrumentation amplifier t opology with four resistors to set the gain. the equation is simply that of a noninverting amplifier as shown in the figure. the two resistors labeled r1 should be closely matched to each other as well as both resistors labeled r2 to ensure acceptable common-mode rejection performance. resistor networks ensure the closest matching as well as matched drifts for good temperature stability. capacitor c1 is included to limit the bandwidth and, therefore, the noise in sensitive applications. the value of this capacitor should be adjusted depending on the desired closed-loop bandwidth of the instrumentation amplifier. the rc combination creates a pole at a frequency equal to 1/(2 r1c1). if the ac-cmrr is critical, then a matched capacitor to c1 should be included across the second resistor labeled r1. because these amplifiers accept rail-to-rail inputs, their input common mode range includes both ground and the positive supply of 1.5v. furthermore, their rail-to-rail output range ensures the widest signal range possible and maximizes the dynamic range of the system. also, with their low supply current of 0.6 a per amplifier, this circuit consumes a quiescent current of only ~1.3 a, yet it still exhibits a 1-khz bandwidth at a circuit gain of 2. driving capacitive loads while the amplifiers? internal gain-bandwidth product is 4khz, both are capable of driving capacitive loads up to 50pf in voltage follower configurations without any additional components. in many applications, however, an operational amplifier is required to drive much larger capacitive loads. the amplifier?s output impedance and a large capacitive load create additional phase lag that further reduces the amplifier?s phase margin. if enough phase delay is introduced, the amplifier?s phase margin is reduced. the effect is quite evident when the transient response is observed as th ere will appear noticeable peaking/ringing in the output transient response. if any amplifier is used in an application that requires driving larger capacitive loads, an isolation resistor between the output and the capacitive load should be used as illustrated in figure 5. table 1 illustrates a range of r iso values as a function of the external c load on the output of these amplifiers. the power supply voltage applied on the these amplifiers at which t hese resistor values were determined empirically wa s 1.8v. the oscilloscope capture shown in figure 6 illustrates a typical transient response obtained with a c load = 500pf and an r iso = 50k ? . note that as c load is increased a smaller r iso is needed for optimal transient response. external capacitive load, c load external output isolation resistor, r iso 0-50pf not required 100pf 120k ? 500pf 50k ? 1nf 33k ? 5nf 18k ? 10nf 13k ? figure 5: using an external resistor to isolate a c load from the amplifer?s output. figure 4: a two op amp instrumentation amplifier.
ts1002/TS1004 page 10 ts1002/4 rev. 1.0 in the event that an external r load in parallel with c load appears in the application, the use of an r iso results in gain accuracy lo ss because the external series r iso forms a voltage-divider with the external load resistor r load . configuring the ts1002 or the TS1004 into a nanowatt analog comparator although optimized for use as an operational amplifier, these amplifiers can also be used as a rail- to-rail i/o comparator as illustrated in figure 7. external hysteresis can be employed to minimize the risk of output oscillati on. the positive feedback circuit causes the input threshold to change when the output voltage changes state. the diagram in figure 8 illustrates the amp lifiers? analog comparator hysteresis band and output transfer characteristic. the design of an analog comparator using the ts1002 or the TS1004 is straightforward. in this application, a 1.5-v power supply (v dd ) was used and the resistor divider network formed by rd1 and rd2 generated a convenient reference voltage (v ref ) for the circuit at ? the supply voltage, or 0.75v, while keeping the current drawn by this resistor divider low. capacito r c1 is used to filter any extraneous noise that could couple into the amplifer?s inverting input. in this application, the desired hysteresis band was set to 100mv (v hyb ) with a desired high trip-point (v hi ) set at 1v and a desired low trip-point (v lo ) set at 0.9v. since these amplifers draw very little supply current (0.6a per amplifier, typica l), it is desired that the design of an analog comparator using these amplfiers should also us e as little current as practical. the first step in the design, therefore, was to set the feedback resistor r3: r3 = 10m ? calculating a value for r1 is given by the following expression: r1 = r3 x (v hyb /v dd ) substituting v hyb = 100mv, v dd = 1.5v, and r3 = 10m ? into the equation above yields: r1 = 667k ? the following expression was then used to calculate a value for r2: r2 = 1/[v hi /(v ref x r1) ? (1/r1) ? (1/r3)] substituting v hi = 1v, v ref = 0.75v, r1 = 667k ? , and r3 = 10m ? into the above expression yields: r2 = 2.5m ? figure 8: analog comparator hysteresis band and output switching points. figure 7: a nanowatt analog comparator with user- programmable hysteresis. figure 6 : ts1002/TS1004 transient response for r iso = 50k ? and c load = 500pf . v in v out
ts1002/TS1004 ts1002/4 rev. 1.0 page 11 printed circuit board layout considerations even though these amplifiers operate from a single 0.65v to 2.5v power supply and consume very little supply current, it is always good engineering practice to bypass the power supplies pins with a 0.1 f ceramic capacitor placed in close proximity to the v dd and v ss (or gnd) pins. good pcb layout techniques and analog ground plane management improve the performance of any analog circuit by decreasing the amount of stray capacitance that could be introduced at the op amp's inputs and outputs. excess stray capacitance can easily couple noise into the input leads of the op amp and excess stray capacitance at the output will add to any external capacitive load. therefore, pc board trace lengths and external component leads should be kept a short as practical to any of the amplifiers? package pins. second, it is also good engineering practice to route/remove any analog ground plane from the inputs and the output pins of these amplifiers.
ts1002/TS1004 page 12 ts1002/4 rev. 1.0 package outline drawing 8-pin msop package outline drawing (n.b., drawings are not to scale)
ts1002/TS1004 silicon laboratories, inc. page 13 400 west cesar chavez, austin, tx 78701 ts1002/4 rev. 1.0 +1 (512) 416-8500 ? www.silabs.com package outline drawing 14-pin tssop package outline drawing (n.b., drawings are not to scale) patent notice silicon labs invests in research and development to help our custom ers differentiate in the market with innovative low-power, s mall size, analog-intensive mixed-signal solutions. s ilicon labs' extensive patent portfolio is a testament to our unique approach and wor ld-class engineering team. the information in this document is believed to be accurate in all respects at the time of publication but is subject to change without notice. silicon laboratories assumes no responsibility for errors and om issions, and disclaims responsib ility for any consequences resu lting from the use of information included herein. additionally, silicon laborat ories assumes no responsibility for the functioning of undescr ibed features or parameters. silicon laboratories reserves the right to make c hanges without further notice. silicon laboratories makes no warra nty, representation or guarantee regarding the suitability of its pr oducts for any particular purpose, nor does silicon laboratories assume any liability arising out of the application or use of any product or circ uit, and specifically disclaims any and all liability, in cluding without limitation consequential or incidental damages. silicon laboratories products are not designed, intended, or authorized for use in applica tions intended to support or sustain life, or for any other application in wh ich the failure of the silicon laboratories product could create a situation where personal injury or death may occur. should buyer purchase or use silicon laboratories products for any such unintended or unaut horized application, buyer shall indemnify and hold silicon laboratories harmless against all claims and damages. silicon laboratories and silicon labs are tr ademarks of silicon laboratories inc. other products or brandnames mentioned herein are trademarks or registered trademarks of their respective holders.
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